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3-1 tm file number 3663.4 caution: these devices are sensitive to electrostatic discharge; follow proper ic handling procedures. 1-888-intersil or 321-724-7143 | intersil and design is a trademark of intersil corporation. | copyright intersil corporation 2000 HFA3101 gilbert cell uhf transistor array the HFA3101 is an all npn transistor array con?ured as a multiplier cell. based on intersils bonded wafer uhf-1 soi process, this array achieves very high f t (10ghz) while maintaining excellent h fe and v be matching characteristics that have been maximized through careful attention to circuit design and layout, making this product ideal for communication circuits. for use in mixer applications, the cell provides high gain and good cancellation of 2nd order distortion terms. pinout HFA3101 (soic) top view features high gain bandwidth product (f t ) . . . . . . . . . . . . . 10ghz high power gain bandwidth product . . . . . . . . . . . . 5ghz current gain (h fe ) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70 low noise figure (transistor). . . . . . . . . . . . . . . . . . 3.5db excellent h fe and v be matching low collector leakage current . . . . . . . . . . . . . . <0.01na pin to pin compatible to upa101 applications balanced mixers multipliers demodulators/modulators automatic gain control circuits phase detectors fiber optic signal processing wireless communication systems wide band ampli?ation stages radio and satellite communications high performance instrumentation ordering information part number (brand) temp. range ( o c) package pkg. no. HFA3101b (h3101b) -40 to 85 8 ld soic m8.15 HFA3101b96 (h3101b) -40 to 85 8 ld soic tape and reel m8.15 1 2 3 4 8 7 6 5 q 5 q 6 q 1 q 2 q 3 q 4 note: q 5 and q 6 - 2 paralleled 3 m x 50 m transistors q 1 , q 2 , q 3 , q 4 - single 3 m x 50 m transistors data sheet september 1998
3-2 absolute maximum ratings thermal information v ceo , collector to emitter voltage . . . . . . . . . . . . . . . . . . . . . . 8.0v v cbo , collector to base voltage . . . . . . . . . . . . . . . . . . . . . . . 12.0v v ebo , emitter to base voltage . . . . . . . . . . . . . . . . . . . . . . . . . 5.5v i c , collector current. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30ma operating conditions temperature range . . . . . . . . . . . . . . . . . . . . . . . . . . -40 o c to 85 o c thermal resistance (typical, note 1) ja ( o c/w) soic package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 185 maximum junction temperature (die) . . . . . . . . . . . . . . . . . . .175 o c maximum junction temperature (plastic package) . . . . . . . . .150 o c maximum storage temperature range . . . . . . . . . . -65 o c to 150 o c maximum lead temperature (soldering 10s) . . . . . . . . . . . . .300 o c (soic - lead tips only) caution: stresses above those listed in ?bsolute maximum ratings may cause permanent damage to the device. this is a stress only rating and operatio n of the device at these or any other conditions above those indicated in the operational sections of this speci?ation is not implied. note: 1. ja is measured with the component mounted on an evaluation pc board in free air. electrical speci?ations t a = 25 o c parameter test conditions (note 2) test level min typ max units collector to base breakdown voltage, v (br)cbo ,q 1 thru q 6 i c = 100 a, i e = 0 a 12 18 - v collector to emitter breakdown voltage, v (br)ceo , q 5 and q 6 i c = 100 a, i b = 0 a 8 12 - v emitter to base breakdown voltage, v (br)ebo , q 1 thru q 6 i e = 10 a, i c = 0 a 5.5 6 - v collector cutoff current, i cbo , q 1 thru q 4 v cb = 8v, i e = 0 a - 0.1 10 na emitter cutoff current, i ebo , q 5 and q 6 v eb = 1v, i c = 0 a - - 200 na dc current gain, h fe , q 1 thru q 6 i c = 10ma, v ce = 3v a 40 70 - collector to base capacitance, c cb q 1 thru q 4 v cb = 5v, f = 1mhz c - 0.300 - pf q 5 and q 6 - 0.600 - pf emitter to base capacitance, c eb q 1 thru q 4 v eb = 0, f = 1mhz b - 0.200 - pf q 5 and q 6 - 0.400 - pf current gain-bandwidth product, f t q 1 thru q 4 i c = 10ma, v ce = 5v c - 10 - ghz q 5 and q 6 i c = 20ma, v ce = 5v c - 10 - ghz power gain-bandwidth product, f max q 1 thru q 4 i c = 10ma, v ce = 5v c - 5 - ghz q 5 and q 6 i c = 20ma, v ce = 5v c - 5 - ghz available gain at minimum noise figure, g nfmin , q 5 and q 6 i c = 5ma, v ce = 3v f = 0.5ghz c - 17.5 - db f = 1.0ghz c - 11.9 - db minimum noise figure, nf min , q 5 and q 6 i c = 5ma, v ce = 3v f = 0.5ghz c - 1.7 - db f = 1.0ghz c - 2.0 - db 50 ? noise figure, nf 50 ? , q 5 and q 6 i c = 5ma, v ce = 3v f = 0.5ghz c - 2.25 - db f = 1.0ghz c - 2.5 - db dc current gain matching, h fe1 /h fe2 , q 1 and q 2 , q 3 and q 4 , and q 5 and q 6 i c = 10ma, v ce = 3v a 0.9 1.0 1.1 input offset voltage, v os , (q 1 and q 2 ), (q 3 and q 4 ), (q 5 and q 6 ) i c = 10ma, v ce = 3v a - 1.5 5 mv input offset current, i c , (q 1 and q 2 ), (q 3 and q 4 ), (q 5 and q 6 ) i c = 10ma, v ce = 3v a - 5 25 a input offset voltage tc, dv os /dt, (q1 and q2, q3 and q 4 , q 5 and q 6 ) i c = 10ma, v ce = 3v c - 0.5 - v/ o c collector to collector leakage, i trench-leakage ? v test = 5v b - 0.01 - na note: 2. test level: a. production tested, b. typical or guaranteed limit based on characterization, c. design typical for information only. HFA3101
3-3 pspice model for a 3 m x 50 m transistor .model nuhfarry npn + (is = 1.840e-16 xti = 3.000e+00 eg = 1.110e+00 vaf = 7.200e+01 + var = 4.500e+00 bf = 1.036e+02 ise = 1.686e-19 ne = 1.400e+00 + ikf = 5.400e-02 xtb = 0.000e+00 br = 1.000e+01 isc = 1.605e-14 + nc = 1.800e+00 ikr = 5.400e-02 rc = 1.140e+01 cjc = 3.980e-13 + mjc = 2.400e-01 vjc = 9.700e-01 fc = 5.000e-01 cje = 2.400e-13 + mje = 5.100e-01 vje = 8.690e-01 tr = 4.000e-09 tf = 10.51e-12 + itf = 3.500e-02 xtf = 2.300e+00 vtf = 3.500e+00 ptf = 0.000e+00 + xcjc = 9.000e-01 cjs = 1.689e-13 vjs = 9.982e-01 mjs = 0.000e+00 + re = 1.848e+00 rb = 5.007e+01 rbm = 1.974e+00 kf = 0.000e+00 + af = 1.000e+00) common emitter s-parameters of 3 m x 50 m transistor freq. (hz) |s 11 | phase(s 11 )|s 12 | phase(s 12 )|s 21 | phase(s 21 )|s 22 | phase(s 22 ) v ce = 5v and i c = 5ma 1.0e+08 0.83 -11.78 1.41e-02 78.88 11.07 168.57 0.97 -11.05 2.0e+08 0.79 -22.82 2.69e-02 68.63 10.51 157.89 0.93 -21.35 3.0e+08 0.73 -32.64 3.75e-02 59.58 9.75 148.44 0.86 -30.44 4.0e+08 0.67 -41.08 4.57e-02 51.90 8.91 140.36 0.79 -38.16 5.0e+08 0.61 -48.23 5.19e-02 45.50 8.10 133.56 0.73 -44.59 6.0e+08 0.55 -54.27 5.65e-02 40.21 7.35 127.88 0.67 -49.93 7.0e+08 0.50 -59.41 6.00e-02 35.82 6.69 123.10 0.62 -54.37 8.0e+08 0.46 -63.81 6.27e-02 32.15 6.11 119.04 0.57 -58.10 9.0e+08 0.42 -67.63 6.47e-02 29.07 5.61 115.57 0.53 -61.25 1.0e+09 0.39 -70.98 6.63e-02 26.45 5.17 112.55 0.50 -63.96 1.1e+09 0.36 -73.95 6.75e-02 24.19 4.79 109.91 0.47 -66.31 1.2e+09 0.34 -76.62 6.85e-02 22.24 4.45 107.57 0.45 -68.37 1.3e+09 0.32 -79.04 6.93e-02 20.53 4.15 105.47 0.43 -70.19 1.4e+09 0.30 -81.25 7.00e-02 19.02 3.89 103.57 0.41 -71.83 1.5e+09 0.28 -83.28 7.05e-02 17.69 3.66 101.84 0.40 -73.31 1.6e+09 0.27 -85.17 7.10e-02 16.49 3.45 100.26 0.39 -74.66 1.7e+09 0.25 -86.92 7.13e-02 15.41 3.27 98.79 0.38 -75.90 1.8e+09 0.24 -88.57 7.17e-02 14.43 3.10 97.43 0.37 -77.05 1.9e+09 0.23 -90.12 7.19e-02 13.54 2.94 96.15 0.36 -78.12 2.0e+09 0.22 -91.59 7.21e-02 12.73 2.80 94.95 0.35 -79.13 2.1e+09 0.21 -92.98 7.23e-02 11.98 2.68 93.81 0.35 -80.09 2.2e+09 0.20 -94.30 7.25e-02 11.29 2.56 92.73 0.34 -80.99 2.3e+09 0.20 -95.57 7.27e-02 10.64 2.45 91.70 0.34 -81.85 2.4e+09 0.19 -96.78 7.28e-02 10.05 2.35 90.72 0.33 -82.68 2.5e+09 0.18 -97.93 7.29e-02 9.49 2.26 89.78 0.33 -83.47 2.6e+09 0.18 -99.05 7.30e-02 8.96 2.18 88.87 0.33 -84.23 2.7e+09 0.17 -100.12 7.31e-02 8.47 2.10 88.00 0.33 -84.97 HFA3101
3-4 2.8e+09 0.17 -101.15 7.31e-02 8.01 2.02 87.15 0.33 -85.68 2.9e+09 0.16 -102.15 7.32e-02 7.57 1.96 86.33 0.33 -86.37 3.0e+09 0.16 -103.11 7.32e-02 7.16 1.89 85.54 0.33 -87.05 v ce = 5v and i c = 10ma 1.0e+08 0.72 -16.43 1.27e-02 75.41 15.12 165.22 0.95 -14.26 2.0e+08 0.67 -31.26 2.34e-02 62.89 13.90 152.04 0.88 -26.95 3.0e+08 0.60 -43.76 3.13e-02 52.58 12.39 141.18 0.79 -37.31 4.0e+08 0.53 -54.00 3.68e-02 44.50 10.92 132.57 0.70 -45.45 5.0e+08 0.47 -62.38 4.05e-02 38.23 9.62 125.78 0.63 -51.77 6.0e+08 0.42 -69.35 4.31e-02 33.34 8.53 120.37 0.57 -56.72 7.0e+08 0.37 -75.26 4.49e-02 29.47 7.62 116.00 0.51 -60.65 8.0e+08 0.34 -80.36 4.63e-02 26.37 6.86 112.39 0.47 -63.85 9.0e+08 0.31 -84.84 4.72e-02 23.84 6.22 109.36 0.44 -66.49 1.0e+09 0.29 -88.83 4.80e-02 21.75 5.69 106.77 0.41 -68.71 1.1e+09 0.27 -92.44 4.86e-02 20.00 5.23 104.51 0.39 -70.62 1.2e+09 0.25 -95.73 4.90e-02 18.52 4.83 102.53 0.37 -72.28 1.3e+09 0.24 -98.75 4.94e-02 17.25 4.49 100.75 0.35 -73.76 1.4e+09 0.22 -101.55 4.97e-02 16.15 4.19 99.16 0.34 -75.08 1.5e+09 0.21 -104.15 4.99e-02 15.19 3.93 97.70 0.33 -76.28 1.6e+09 0.20 -106.57 5.01e-02 14.34 3.70 96.36 0.32 -77.38 1.7e+09 0.20 -108.85 5.03e-02 13.60 3.49 95.12 0.31 -78.41 1.8e+09 0.19 -110.98 5.05e-02 12.94 3.30 93.96 0.31 -79.37 1.9e+09 0.18 -113.00 5.06e-02 12.34 3.13 92.87 0.30 -80.27 2.0e+09 0.18 -114.90 5.07e-02 11.81 2.98 91.85 0.30 -81.13 2.1e+09 0.17 -116.69 5.08e-02 11.33 2.84 90.87 0.30 -81.95 2.2e+09 0.17 -118.39 5.09e-02 10.89 2.72 89.94 0.29 -82.74 2.3e+09 0.16 -120.01 5.10e-02 10.50 2.60 89.06 0.29 -83.50 2.4e+09 0.16 -121.54 5.11e-02 10.13 2.49 88.21 0.29 -84.24 2.5e+09 0.16 -122.99 5.12e-02 9.80 2.39 87.39 0.29 -84.95 2.6e+09 0.15 -124.37 5.12e-02 9.49 2.30 86.60 0.29 -85.64 2.7e+09 0.15 -125.69 5.13e-02 9.21 2.22 85.83 0.29 -86.32 2.8e+09 0.15 -126.94 5.13e-02 8.95 2.14 85.09 0.29 -86.98 2.9e+09 0.15 -128.14 5.14e-02 8.71 2.06 84.36 0.29 -87.62 3.0e+09 0.14 -129.27 5.15e-02 8.49 1.99 83.66 0.29 -88.25 common emitter s-parameters of 3 m x 50 m transistor (continued) freq. (hz) |s 11 | phase(s 11 )|s 12 | phase(s 12 )|s 21 | phase(s 21 )|s 22 | phase(s 22 ) HFA3101
3-5 application information the HFA3101 array is a very versatile rf building block. it has been carefully laid out to improve its matching properties, bringing the distortion due to area mismatches, thermal distribution, betas and ohmic resistances to a minimum. the cell is equivalent to two differential stages built as two ?ariable transconductance multipliers?in parallel, with their outputs cross coupled. this con?uration is well known in the industry as a gilbert cell which enables a four quadrant multiplication operation. due to the input dynamic range restrictions for the input levels at the upper quad transistors and lower tail transistors, the HFA3101 cell has restricted use as a linear four quadrant multiplier. however, its con?uration is well suited for uses where its linear response is limited to one of the inputs only, as in modulators or mixer circuit applications. examples of these circuits are up converters, down converters, frequency doublers and frequency/phase detectors. although linearization is still an issue for the lower pair input, emitter degeneration can be used to improve the dynamic range and consequent linearity. the HFA3101 has the lower pair emitters brought to external pins for this purpose. in modulators applications, the upper quad transistors are used in a switching mode where the pairs q 1 /q 2 and q 3 /q 4 act as non saturating high speed switches. these switches are controlled by the signal often referred as the carrier input. the signal driving the lower pair q 5 /q 6 is commonly used as the modulating input. this signal can be linearly transferred to the output by either the use of low signal levels (well below the thermal voltage of 26mv) or by the use of emitter degeneration. the chopped waveform appearing at the output of the upper pair (q 1 to q 4 ) resembles a signal that is multiplied by +1 or -1 at every half cycle of the switching waveform. figure 1 shows the typical input waveforms where the frequency of the carrier is higher than the modulating signal. the output waveform shows a typical suppressed carrier output of an up converter or an am signal generator. carrier suppression capability is a property of the well known balanced modulator in which the output must be zero when one or the other input (carrier or modulating signal) is equal to zero. however, at very high frequencies, high frequency mismatches and ac offsets are always present and the suppression capability is often degraded causing carrier and modulating feedthrough to be present. being a frequency translation circuit, the balanced modulator has the properties of translating the modulating frequency ( m ) to the carrier frequency ( c ), generating the two side bands u = c + m and l = c - m . figure 2 shows some translating schemes being used by balanced mixers. carrier signal modulating signal differential output +1 -1 figure 1. typical modulator signals figure 2a. up conversion or suppressed carrier am figure 2b. down conversion figure 2c. zero if or direct down conversion figure 2. modulator frequency spectrum c + m c - m c if ( c - m ) folded back m c baseband c m HFA3101
3-6 the use of the HFA3101 as modulators has several advantages when compared to its counterpart, the diode doublebalanced mixer, in which it is required to receive enough energy to drive the diodes into a switching mode and has also some requirements depending on the frequency range desired, of different transformers to suit speci? frequency responses. the HFA3101 requires very low driving capabilities for its carrier input and its frequency response is limited by the f t of the devices, the design and the layout techniques being utilized. up conversion uses, for uhf transmitters for example, can be performed by injecting a modulating input in the range of 45mhz to 130mhz that carries the information often called if (intermediate frequency) for up conversion (the if signal has been previously modulated by some modulation scheme from a baseband signal of audio or digital information) and by injecting the signal of a local oscillator of a much higher frequency range from 600mhz to 1.2ghz into the carrier input. using the example of a 850mhz carrier input and a 70mhz if, the output spectrum will contain a upper side band of 920mhz, a lower side band of 780mhz and some of the carrier (850mhz) and if (70mhz) feedthrough. a band pass filter at the output can attenuate the undesirable signals and the 920mhz signal can be routed to a transmitter rf power amplifier. down conversion, as the name implies, is the process used to translate a higher frequency signal to a lower frequency range conserving the modulation information contained in the higher frequency signal. one very common typical down conversion use for example, is for superheterodyne radio receivers where a translated lower frequency often referred as intermediate frequency (if) is used for detection or demodulation of the baseband signal. other application uses include down conversion for special ?tering using frequency translation methods. an oscillator referred as the local oscillator (lo) drives the upper quad transistors of the cell with a frequency called c . the lower pair is driven by the rf signal of frequency m to be translated to a lower frequency if. the spectrum of the if output will contain the sum and difference of the frequencies c and m . notice that the difference can become negative when the frequency of the local oscillator is lower than the incoming frequency and the signal is folded back as in figure 2. note: the acronyms rf, if and lo are often interchanged in the industry depending on the application of the cell as mixers or modulators. the output of the cell also contains multiples of the frequency of the signal being fed to the upper quad pair of transistors because of the switching action equivalent to a square wave multiplication. in practice, however, not only the odd multiples in the case of a symmetrical square wave but some of the even multiples will also appear at the output spectrum due to the nature of the actual switching waveform and high frequency performance. by-products of the form m* c +n* m with m and n being positive or negative integers are also expected to be present at the output and their levels are carefully examined and minimized by the design. this distortion is considered one of the ?ures of merit for a mixer application. the process of frequency doubling is also understood by having the same signal being fed to both modulating and carrier ports. the output frequency will be the sum of c and m which is equivalent to the product of the input frequency by 2 and a zero hz or dc frequency equivalent to the difference of c and m . figure 2 also shows one technique in use today where a process of down conversion named zero if is made by using a local oscillator with a very pure signal frequency equal to the incoming rf frequency signal that contains a baseband (audio or digital signal) modulation. although complex, the extraction or detection of the signal is straightforward. another useful application of the HFA3101 is its use as a high frequency phase detector where the two signals are fed to the carrier and modulation ports and the dc information is extracted from its output. in this case, both ports are utilized in a switching mode or overdrive, such that the process of multiplication takes place in a quasi digital form (2 square waves). one application of a phase detector is frequency or phase demodulation where the fm signal is split before the modulating and carrier ports. the lower input port is always 90 degrees apart from the carrier input signal through a high q tuned phase shift network. the network, being tuned for a precise 90 degrees shift at a nominal frequency, will set the two signals 90 degrees apart and a quiescent output dc level will be present at the output. when the input signal is frequency modulated, the phase shift of the signal coming from the network will deviate from 90 degrees proportional to the frequency deviation of the fm signal and a dc variation at the output will take place, resembling the demodulated fm signal. the HFA3101 could also be used for quadrature detection, (i/q demodulation), agc control with limited range, low level multiplication to name a few other applications. biasing various biasing schemes can be employed for use with the HFA3101. figure 3 shows the most common schemes. the biasing method is a choice of the designer when cost, thermal dependence, voltage overheads and dc balancing properties are taken into consideration. figure 3a shows the simplest form of biasing the HFA3101. the current source required for the lower pair is set by the voltage across the resistor r bias less a v be drop of the lower transistor. to increase the overhead, collector resistors are substituted by an rf choke as the upper pair functions as a current source for ac signals. the bases of the upper and lower transistors are biased by r b1 and r b2 respectively. the voltage drop across the resistor r 2 must be higher than a v be with an increase suf?ient to assure that the collector to base junctions of the lower pair are always reverse biased. notice that this same voltage also sets the v ce of operation of the lower pair which is important for the optimization of gain. resistors r ee are nominally zero for applications where the input signals are well below 25mv peak. resistors r ee are used to increase the linearity HFA3101
3-7 of the circuit upon higher level signals. the drop across r ee must be taken into consideration when setting the current source value. figure 3b depicts the use of a common resistor sharing the current through the cell which is used for temperature compensation as the lower pair v be drop at the rate of -2mv/ o c. figure 3c uses a split supply. design example: down converter mixer figure 4 shows an example of a low cost mixer for cellular applications. the design ?xibility of the HFA3101 is demonstrated by a low cost, and low voltage mixer application at the 900mhz range. the choice of good quality chip components with their self resonance outside the boundaries of the application are important. the design has been optimized to accommodate the evaluation of the same layout for various quiescent current values and lower supply voltages. the choice of r e became important for the available overhead and also for maintaining an ac true impedance for high frequency signals. the value of 27 ? has been found to be the optimum minimum for the application. the input impedances of the HFA3101 base input ports are high enough to permit their termination with 50 ? resistors. notice the ac termination by decoupling the bias circuit through good quality capacitors. the choice of the bias has been related to the available power supply voltage with the values of r 1 , r 2 and r bias splitting the voltages for optimum v ce values. for evaluation of the cell quiescent currents, the voltage at the emitter resistor r e has been recorded. the gain of the circuit, being a function of the load and the combined emitter resistances at high frequencies have been kept to a maximum by the use of an output match network. the high output impedance of the HFA3101 permits figure 3a. figure 3b. figure 3c. figure 3. v cc r b1 r 1 r 2 r bias r e r ee r ee l ch 1 2 3 4 8 7 6 5 q 5 q 6 q 1 q 2 q 3 q 4 r b2 v cc r b1 r 1 r 2 r bias r e r ee r ee 1 2 3 4 8 7 6 5 q 5 q 6 q 1 q 2 q 3 q 4 r b2 r c l ch v ee r b1 r 1 r bias r e r ee r ee 1 2 3 4 8 7 6 5 q 5 q 6 q 1 q 2 q 3 q 4 r b2 v cc l ch r 2 27 l ch 1 2 3 4 8 7 6 5 q 5 q 6 q 1 q 2 q 3 q 4 v cc 390nh 0.01 0.01 110 220 0.1 v cc 3v 75mhz 2k 5p to 12p lo in 51 825mhz 51 900mhz if out rf in 0.01 0.01 0.01 330 figure 4. 3v down converter application HFA3101
3-8 broadband match if so desired at 50 ? (r l =50 ? to 2k ? )as well as with tuned medium q matching networks (l, t etc.). stability the cell, by its nature, has very high gain and precautions must be taken to account for the combination of signal re?ctions, gain, layout and package parasitics. the rule of thumb of avoiding re?cted waves must be observed. it is important to assure good matching between the mixer stage and its front end. laboratory measurements have shown some susceptibility for oscillation at the upper quad transistors input. any lo pre?tering has to be designed such the return loss is maintained within acceptable limits specially at high frequencies. typical off the shelf ?ters exhibits very poor return loss for signals outside the passband. it is suggested that a ?ad?or a broadband resistive network be used to interface the lo port with a ?ter. the inclusion of a parallel 2k resistor in the load decreases the gain slightly which improves the stability factor and also improves the distortion products (output intermodulation or 3rd order intercept). the employment of good rf techniques shall suf?e the stability requirements. evaluation the evaluation of the HFA3101 in a mixer con?uration is presented in figures 6 to 11, table 1 and table 2. the layout is depicted in figure 5. the output matching network has been designed from data taken at the output port at various test frequencies with the setup as in table 1. s 22 characterization is enough to assure the calculation of l, t or transmission line matching networks. figure 5. up/down converter layout, 400%; material g10, 0.031 table 1. s 22 parameters for down conversion, l ch = 10 h frequency resistance reactance 10mhz 265 ? 615 ? 45mhz 420 ? - 735 ? 75mhz 122 ? - 432 ? 100mhz 67 ? - 320 ? table 2. typical parameters for down conversion, l ch = 10 h parameter lo level v cc = 3v, i bias = 8ma power gain -6dbm 8.5db toi output -6dbm 11.5dbm nf ssb -6dbm 14.5db power gain 0dbm 8.6db toi output 0dbm 11dbm nf ssb 0dbm 15db parameter lo level v cc = 4v, i bias = 19ma power gain -6dbm 10db toi output -6dbm 13dbm nf ssb -6dbm 20db power gain 0dbm 11db toi output 0dbm 12.5dbm nf ssb 0dbm 24db table 3. t ypical values of s 22 for the output port. l ch = 390nh i bias = 8ma (set up of figure 11) frequency resistance reactance 300mhz 22 ? -115 ? 600mhz 7.5 ? -43 ? 900mhz 5.2 ? -14 ? 1.1ghz 3.9 ? 0 ? table 4. typical values of s22. l ch = 390nh, i bias = 18ma frequency resistance reactance 300mhz 23.5 ? -110 ? 600mhz 10.3 ? -39 ? 900mhz 8.7 ? -14 ? 1.1ghz 8 ? 0 ? HFA3101
3-9 up converter example an application for a up converter as well as a frequency multiplier can be demonstrated using the same layout, with an addition of matching components. the output port s 22 must be characterized for proper matching procedures and depending on the frequency desired for the output, transmission line transformations can be designed. the return loss of the input ports maintain acceptable values in excess of 1.2ghz which can permit the evaluation of a frequency doubler to 2.4ghz if so desired. the addition of the resistors r ee can increase considerably the dynamic range of the up converter as demonstrated at figure 13. the evaluation results depicted in table 5 have been obtained by a triple stub tuner as a matching network for the output due to the layout constraints. based on the evaluation results it is clear that the cell requires a higher bias current for overall performance. figure 6. output port s 22 test set up figure 7. lo port return loss figure 8. rf port return loss figure 9. if port return loss, with matching network figure 10. typical in band output spectrum, v cc = 3v figure 11. typical out of band output spectrum v cc 3v 0.1 l ch 1 2 3 4 8 7 6 5 q 5 q 6 q 1 q 2 q 3 q 4 2k s 11 0db 5db/div 100mhz 1.1ghz log mag 3v 4v 0db 10db/div 100mhz 1.1ghz s 11 log mag 0db 5db/div 10mhz s 22 log mag 110mhz 76mhz 64m 11 * lo - 10rf 88m 12rf - 13lo if span 40mhz lo = 825mhz -6dbm rf = 901mhz - 25dbm -17dbm 10db/ div 675 750 825 900 975 10db/ div lo + 2rf span 500mhz lo - 2rf -26dbm -36dbm -58dbm -53dbm lo = 825mhz -6dbm rf = 900mhz -25dbm HFA3101
3-10 design example: up converter mixer figure 12 shows an example of an up converter for cellular applications. conclusion the HFA3101 offers the designer a number of choices and different applications as a powerful rf building block. although isolation is degraded from the theoretical results for the cell due to the unbalanced, nondifferential input schemes being used, a number of advantages can be taken into consideration like cost, ?xibility, low power and small outline when deciding for a design. table 5. typical parameters for the up converter example parameter v cc = 3v, i bias = 8ma v cc = 4v, i bias = 18ma power gain, lo = -6dbm 3db 5.5dbm power gain, lo = 0dbm 4db 7.2db rf isolation, lo = 0dbm 15dbc 22dbc lo isolation, lo = 0dbm 28dbc 28dbc figure 12. up converter figure 13. typical spectrum performance of up converter rf in 0.01 390nh 900mhz 5.2nh v cc 3v 0.1 1 2 3 4 8 7 6 5 q 5 q 6 q 1 q 2 q 3 q 4 11p 0.01 75mhz 27 220 r ee r ee 51 lo in v cc 0.01 0.01 110 330 3v 825mhz 0.01 0.01 51 47-100pf 901 912 890 2lo - 10rf 12rf output without emitter degeneration rf = 76mhz lo = 825mhz span 50mhz output with emitter degeneration r ee = 4.7 ? 825 900 976 expanded spectrum r ee = 4.7 ? HFA3101
3-11 typical performance curves for transistors figure 14. i c vs v ce figure 15. h fe vs i c figure 16. gummel plot figure 17. f t vs i c figure 18. gain and noise figure vs frequency note: figures 14 through 18 are only for q 5 and q 6 . v ce (v) i c (ma) 0 2.0 6.0 4.0 0 70 60 50 40 30 20 10 i b = 800 a i b = 1ma i b = 200 a i b = 400 a i b = 600 a h fe i c (a) 10 -10 10 -8 10 -6 10 -4 10 -2 10 0 140 120 100 80 60 40 20 0 v ce = 5v v be (v) i c and i b (a) 10 -10 10 -8 10 -6 10 -4 10 -2 10 0 10 -12 0.20 0.40 0.60 0.80 1.0 v ce = 3v i c (a) f t (ghz) 12 10 8 6 4 2 0 10 -4 10 -3 10 -2 10 -1 20 18 16 14 12 10 4 6 noise figure (db) frequency (ghz) |s 21 | (db) 0.5 1.5 1.0 2.0 0 2.5 3.0 4.8 4.6 4.4 4.2 4.0 3.8 3.6 3.4 3.2 8 HFA3101
3-12 all intersil semiconductor products are manufactured, assembled and tested under iso9000 quality systems certi?ation. intersil semiconductor products are sold by description only. intersil corporation reserves the right to make changes in circuit design and/or spec ifications at any time with- out notice. accordingly, the reader is cautioned to verify that data sheets are current before placing orders. information furnished by intersil is b elieved to be accurate and reliable. however, no responsibility is assumed by intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of th ird parties which may result from its use. no license is granted by implication or otherwise under any patent or patent rights of intersil or its subsidiari es. for information regarding intersil corporation and its products, see web site www.intersil.com die characteristics process uhf-1 die dimensions: 53 mils x 52 mils x 14 mils 1340 m x 1320 m x 355.6 m metallization: type: metal 1: alcu(2%)/tiw thickness: metal 1: 8k ? 0.5k ? type: metal 2: alcu(2%) thickness: metal 2: 16k ? 0.8k ? passivation: type: nitride thickness: 4k ? 0.5k ? substrate potential (powered up): floating metallization mask layout HFA3101 1 1 2233 4 4 5 5 6 6 7 7 8 8 HFA3101

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